VHF Stability
rev AM April 15/12
rev PM Sept 23/11
HF Amplifier
Stability at VHF
While most areas of troubleshooting and engineering follow logical steps, a great deal of
empirical
or voodoo engineering
surrounds
amplifier instability. Few builders follow logical planned steps to understand,
test for,
and correct instability
problems. Much of
the problem surrounds the abstract nature of flaws causing instability, lack of
visualization of component and layout VHF behavior, and the
lack of good articles about causes of instability.
The absence of
readily available
information
creates a vacuum, and that vacuum causes builders to
rely on some well-written but,
regretfully,
pathological
science. When we couple the natural human desire to have a fast, simple, universal answer
or simple instruction to solve
every complex problem, we
become targets for some very bad science.
VHF Impedances
The intelligent discussion of VHF systems requires a feel for VHF systems,
and how VHF energy moves through a system. At VHF, wavelength is very short.
Wavelength in feet is F/983.6, a typical 150 MHz system would have a quarter
wavelength of 19.8 inches. This does not include velocity factor of dielectric,
or unevenly distributed series inductances or shunt capacitances, which can make
electrical distance along a conductor appear much longer than it is.
The general rule of thumb is two electrical degree length paths will have a
negligible effect on system impedances. While that is 3 feet on 160 meters, two
electrical degrees is roughly 1/2 inch on 150 MHz.
A 10-inch long conductor, in particular a thin conductor with a dielectric,
is just like having no path at all for VHF, yet we see Internet suggestions of
adding thin wires from grid pins up to tuning capacitors to beneficially alter
the path from tuning capacitor to grid in some amplifier layouts! This is the
same false notion Johnson engineers used in the
Valiant and Ranger
transmitters. In the image below, Johnson used a buss wire to ground tuning
capacitors to the 6146 socket ground, which turns out to be a disaster for
ground loops and VHF harmonic suppression.
A similar fallacy exists in HF amplifiers, where people add an even thinner
and longer wire from tuning capacitors to the grid pins of tube sockets. A
Kenwood TL922 I worked on
had just such a mod, clearly the installer had no idea about wavelengths and
transmission line behavior. They used a thin wire several inches long, shown
below, in an attempt to reduce tuning capacitor to grid path impedance!
VHF paths must be short and very wide, and ideally would be smooth
surfaces. A wide path acts more like a groundplane, instead of a transmission
line. For example, a 20-inch metal radius makes a very low impedance ground path
at VHF, yet a 20-inch thin wire can look like no ground connection at all
between two points! If we want reasonable results, we have to stop following
unreasonable logic or junk science when making changes. If a tuning capacitor is
poorly connected back to the grid at VHF, because the path through 10-inch wide
sheet is too long, we are NOT going to improve it with an additional several
inches length of .060 inch wide conductor in parallel. The notion something so
thin in parallel with a wide ground plane helps reduce path impedance is silly.
In the TL922, I used the chassis groundplane effect to advantage by adding a
wide strap connection only a few inches long.
To actually improve things, we must stop treating VHF systems like they were
DC systems, or HF systems.
A similar error occurs in a west coast amateur’s discussions of VHF
suppressors. He treats the VHF suppressor as an isolated component that solely
determines anode system Q. The anode suppressor is actually one small section of
a much longer path that behaves like a transmission line. The suppressor Q, in
isolation, means very little to overall system behavior. We have to look at the
suppressor in full context of how it modifies a much more complex system’s
overall impedance. This is why every suppressor, when optimized, must be
optimized for a particular system. As we see when we look at commercial designs,
one size does not fit all. In many cases no suppressor at all is required, and
when required, depending on application and layout, a variety of styles are
used. There is no single right way and wrong way.
Circuit Configuration
We might assume, because of a certain circuit configuration or descriptive
system name, the system
behaves like that configuration on all frequencies. More often than not, there
are exceptions to that assumption. Within a certain frequency range, things do behave as schematics
and descriptions might
indicate. Beyond those frequency limits things change. Let’s consider the
case of a grounded grid amplifier.
A grounded grid amplifier only acts as if the grid were grounded over a
certain range. The range where the system behaves as expected is determined by
how the grid is grounded, both with components and wiring the builder or
designer can control, and by things inside the tube that only the tube
manufacturer can control. Grounded grid amplifiers are theoretically
unconditionally stable, because they have extremely high negative feedback from
output-to-input. This negative feedback, determined by the ratio of output load
impedance to input impedance, normally swaps out any regeneration. The grid or
grids, in theory anyway, also shield the input from the output.
Unexpected problems occur when the system does
not behave as the schematic implies. This is because the
schematic does not show stray impedances. These stray impedances can reconfigure
the amplifier stage on various frequencies, making it change modes anywhere from stable
grounded-grid operation, to a mode where the grid floats.
3-500Z tube capacitances. Dominant capacitance is
grid-anode and grid-cathode.
These capacitance ratios are loosely typical for most transmitting tubes. The
grid-to-chassis impedance is why the
anode-grid path, not the plate-cathode path, dominates VHF stability.
At HF and lower, a different effect can occur. The plate-cathode feedthrough
can allow regeneration from anode to cathode. At high frequencies with long thin
grid leads, the might not appear grounded at all.
This mixture of effects varies with input and output networks, layout, tube,
socket, and wiring.
For the 8877, published capacitances are:
The large area grid, with
very close spacing
to the cathode, causes abnormally high cathode-to-grid capacitance. This
capacitance actually helps stability, because it holds the grid closer to
cathode potential.
The dominant reason the 8877 is so stable is the cone-type grid support
structure that exits the envelope through a large area flange. If the grid
flange is directly grounded with a very short chassis ground, grid impedance is
so low that the 8877 is unconditionally stable without any suppression at all.
The very same things that allow a tube to be a good amplifier tube on VHF and
UHF make it a stable tube at HF. The least stable tubes are those with poorest
VHF and UHF performance. If a tube is useful and stable at VHF or UHF, it will
be even more stable in a properly laid-out HF amplifier system.
When the impedance of Lgs becomes large enough, either resistive, inductive,
or capacitive, the tube is effectively no longer in grounded grid operation. We
now have the makings of a TPTG oscillator, with feedback through Cpg.
This is why it is important to ground the grid as well as possible. Look at
the changes I made in a TL922
grid system. The same applies to the SB220.
Cause of Unwanted VHF
parasitic Oscillations
The most sensitive
control element in
the tube (the control grid) generally
has the largest
influence in
determining
oscillation
frequency. The grid often controls if, when, and where the system
oscillates. Small changes in control grid (and screen grid) voltage, with
respect to cathode, dominant tube operation. This is
why primary
frequency control
elements of VFO’s or
crystal oscillators
are normally placed in the control grid’s
circuitry.
The
anode system
normally has the
highest RF voltage
swing in the system. The anode-cathode path through the tube has the largest
time-varying current, the current being primarily regulated by the control grid. If the anode system presents a
high impedance at
radio frequencies, small changes in anode current will cause significant RF
voltages to appear. With large anode voltage changes for relatively small anode
current changes, a very small amount of anode-to-grid capacitance can be enough to form
an oscillator. Logically, the grid-anode path and circuitry at the grid and
anode by far
are most likely
to dominate an unwanted oscillator system.
An oscillator also must have enough gain
to overcome feedback loss, and feedback has to be the correct additive phase.
If positive or
regenerative feedback does not exceed system
losses, the system
cannot oscillate. The
control grid-to-anode path generally
has the highest
possible unwanted gain in the
amplifier system,
and that is why this
part of the system
is (by far) the most
problematic area for unwanted VHF stability problems in lower frequency
amplifiers. The
normal mode of VHF
oscillation in HF
PA’s is where the
tube becomes a
tuned-plate tuned-grid oscillator. The frequency of this oscillator is mostly
determined by the grid system, from the grid inside the tube, out through the
grid terminal, to whatever is outside the tube.
Inside the Tube
The control grid
system has considerable stray capacitance to the cathode and other low
VHF-impedance elements. The control grid also has a conductive path connecting
the grid to the socket grid pin. The combination of grid-to-ground shunt
capacitance and grid-to-ground series inductance through the grid
lead-to-chassis path forms a parallel resonant circuit with fairly high Q. Since
this circuit mostly exists inside the tube, there is very little we can do
externally to reduce grid impedance at very high frequencies. Most of that
impedance is inherent in tube construction. Every grid, deep inside the tube, behaves like
it is connected
to the grid pin or grid ring through a
parallel-tuned
circuit. At some frequency, internal grid stray-capacitance
parallel-tunes
the total inductance
of the grid-to-ground conductor path.
L1 and C2 primary
determine optimum frequency for unwanted oscillations. The parallel
resonant combination
of L1 and C2 “float” the grid
off the chassis.
Unfortunately we
cannot greatly
affect L1-C2,
they are mostly
inside the tube. Optimally, L1/C2 should be
resonant as far
above the highest desired working
frequency as
possible. L1 should have the lowest
possible impedance
below the operating
frequency.
L3/C4
(including the C3+C6
path and C5 path)
allows the anode to
easily change anode voltage at VHF with small current changes inside
the tube. We want
L3/C4 to be resonant
below the grid resonant frequency, with the
lowest possible
reactance. Ideally
we would want the anode to
see zero impedance
at the frequency
where the grid is
resonant, but that is impossible. The next best choice is to load the anode with
a perfect termination, like a dummy load, that has
low-to-modest resistance to ground.
For maximum
operating efficiency we want
L3 to have minimum series
resistance at the
desired operating frequency.
The unwanted VHF feedback
path, creating an undesired oscillator is through C1.
Anode system path
The anode has stray capacitance to chassis and to grid
The long RF connection
from anode-to-chassis has series inductance. This is the anode’s “VHF tank coil”
Stray capacitance from anode-to-ground parallel-tunes the anode wiring’s
inductance
This forms a complex resonant circuit with the anode components and anode
capacitance
Let’s measure a typical amplifier and see what impedance the tube looks into
when the tank is tuned and terminated in 50 ohms.
On twenty meters, the anode system of the above amplifier has at least 5
resonances between 500 kHz and 500 MHz! This is typical for almost any HF
amplifier using a physically large tank system, especially with large
air-variable capacitors. The highest impedances appear in the broadcast band,
and at the operating frequency (right side of Smith chart). Lowest impedances
are at low-order harmonics. This is necessary for harmonic suppression.
Some people claim they can put a grid dip meter near the anode lead and tell
us everything about the Q and frequency of resonances. This is clearly untrue,
because they also falsely claim the resonant frequencies never change as the
band or tuning are changed.
The marker is set on the only frequency range where, when the parasitic
suppressor is removed, this AL80B oscillates.
In the AL80B, on twenty meters, the anode looks into an impedance like this:
Note, at the frequency where a 3-500Z typically oscillates, impedance is 23.4
-j25.6 ohms. This is a Q of 0.9
Despite this low Q, some people suggest “low-Q” nichrome suppressors are the
only fix for all problems. Adding a nichrome suppressor to the AL80B actually
increases VHF Q, while decreasing desirable HF Q of the anode system.
Avoid magical gimmick systems, you are just likely to make things worse!
Neither Eimac, nor any large scale experienced amplifier component or
equipment manufacturers in the world, suggest a reduction of HF Q is necessary
or desirable.
On 15 meters, the impedance changes, even at VHF!
Notice how the trace near the 184 MHz marker, as well as all over the chart,
has now changed. The AL80B now has six resonances.
Again, multiple variable resonances are typical for most HF amplifiers. They
are not in the least harmful, and are actually unavoidable. They have little to
no significance in operation. The only requirement is the amplifier have
adequate harmonic suppression, and pass stability tests.
This is why the PLATE and LOAD, as well as the BAND switch, must be rotated
through all positions to verify amplifier stability.
My stability testing
procedure does just that.
Grid system path
The 3-500Z control grid is a moderately good design. It has three grid pins
(all three should be grounded), a wide grid support, and about 2″ of total
height above the glass base. The outer cage is the grid, and the inner spirals
are the filament helices. The upper end of the filament helices weld to a
vertical rigid support rod:
The grid has inductance in the connection path from grid-to-chassis. This is
the grid circuit’s “VHF tank coil”
The grid has
stray capacitance to chassis and filament. This parallel tunes the grid’s
inductance
The grid also has significant internal capacitance to the
anode which forms the unwanted oscillator’s primary feedback
path
With all of this,
the circuit has
everything needed to
become a tuned-plate
tuned-grid
oscillator if tuning and feedback conditions are right.
If feedback loss
(attenuation) from
anode-to-grid capacitance is
less than tube gain
at some
frequency, the tube may
oscillate. The final
requirement is the
phase of unwanted
feedback must be an angle
value that produces
regenerative or
positive feedback.
These requirements
are the same in any
oscillator.
The cathode drive system is normally not part of VHF instability. With a
proper input system, the cathode system has low enough impedance to be
considered “grounded”.
The suppressor, Rs and L1, in combination with the existing anode path Z,
becomes the anode-to-chassis path impedance. This forms the plate load
resistance at every frequency. The impedance must be analyzed at every
frequency.
The grid path impedance can also be varied, and again must be analyzed at
every frequency of concern.
In a grounded-grid amplifier, grid Z should be as low as possible at every
frequency.
For maximum stability we want the resonant frequency of the anode, at every
frequency where there is significant internal tube gain, to be lower than the
resonant frequency of the control grid.
Conditions for Instability
Once again, the
conditions required
for instability are:
- Gain from grid to plate must
exceed
attenuation or loss in
the feedback
capacitance path from plate-to-grid - The grid must
have a
sufficiently
high impedance
for the amount
of available
feedback to
cause a
stability
problem - The anode must
have a
sufficiently
high impedance
near the same
frequency as
grid resonance to cause instability - Feedback phase
must be correct
to provide positive
or additive voltage
feedback. This best occurs when the plate is resonant near or above the
grid’s parallel resonant frequency
If any one of
the above four
requirements are not
met, the tube will
not oscillate! This
is true no matter
how high Q is
in any individual
path, or if the tube
has suppressors or
not. What this means is, we cannot just look at “Q”. The problem, while we
would like it to be simple with only one possible cause and one universal
solution, really involves four distinct but easily understood areas. This is the
same with any oscillator, whether the oscillator is desired or unintentional!
The Myth of Parasitics Causing Bandswitch and Tube Failures
Claims have been
made that tubes will
remain stable for
years, and a
“sudden
event” (like a photon striking a tube) will
make the tube break
into an uncontrolled
oscillation. Oddly, these claims all come from one source with nothing but the
fact “something bad happened” as evidence of parasitics. His evidence
of a cure is someone installed his kit and was happy. This is like the
tactic Gotham used for selling antennas.
We all know, or hopefully we understand, that oscillators are oscillators.
Oscillations that start cannot stop unless one
or more of the four
important system
parameters above
significantly change.
Also, an uncontrolled oscillation cannot suddenly start, especially one that has
so much feedback it reaches catastrophic levels, unless all of the above
conditions are met. These conditions are either met, or they are not met, unless
we change something. The notion a healthy system can go along for hours, weeks,
or years and suddenly break into an uncontrolled oscillation that damages
components is highly unlikely unless a major component significantly changes
characteristics.
Conversely, if the system is stable, one or more of the above parameters must
change in a way that allows oscillation. If that happens the tube will oscillate
continuously until operating voltages are removed. We can actually intentionally
try to create optimum oscillation conditions. I do that to
test for stability. Even
creating intentional oscillations, nearly all of the time,
oscillations are not
damaging.
Exploding VFO’s Anyone?
Consider
the oscillator in a
transmitter. The
oscillator rapidly
comes up to a state
of equilibrium and
stops increasing in
amplitude. We never
find an oscillator
that can output more
power than the same
tube can
provide operated as
an amplifier, as a matter of fact power is always less!
Any claim an amplifier tube
that saturates at a
few amperes
cathode current can
provide 50 or more
amperes of
“big-bang” current
from accidental oscillation is profoundly
ridiculous. The
cathode can’t
magically produce
more current as an
oscillator than the
saturated emission
would permit in any
other service. Such
big-bang claims
might make good
fictional theater,
but they aren’t
factual.
The most
common effect of unwanted VHF
oscillations are
creation of spurious
signals and odd meter readings; not
bangs, pops, or
arced band switches.
Bangs and pops are
caused by gassy
tubes or other
problems, while arced
bandswitches (if
caused by an
oscillation) are
generally caused by
oscillations at or
near the desired
operating frequency!
Location of
Suppressor
Most of our
modern PA’s are
grounded grid
(cathode driven).
Cathode driven
operation requires
that one or more grids be
directly grounded to
the chassis (at
least for RF) with
the lowest impedance
possible. This is
necessary to shield
the output from the
input, and assure
operating frequency
stability and purity
of emissions.
A VHF
oscillation, if it
happens to occur in
an HF PA, is almost
always rooted in the
system behaving like
a
“tuned-plate/tuned-grid”
oscillator. To be most effective, a VHF suppressor
must be located
between the tube
element and a
low-impedance path
to ground at VHF.
This allows the best loss-loading of
the unwanted
internal oscillator circuits.
The actual working
circuit causing a
VHF oscillation is
almost always
entirely different
than what appears on
the actual
component-based
schematic. The
cathode, an element
commonly involved in
low-frequency
instability is
rarely involved in
VHF oscillations,
other than supplying
electrons and stray
capacitance to
ground.
Even though the anode is
the
second most
problematic area,
it is an area most easily
altered and modified.
This is because virtually every anode has much shorter and wider connection
paths to outside tube terminals. Suppressors are
normally found in
anode systems, even
though other
locations can work
to suppress
oscillations.
Locating the r suppression
in the anode path generally works best
because the grid or
grids can remain
well-grounded for
RF, provided the
best operating
frequency
performance, and because the anode connector is often the shortest access point to tube
internals.
To be most effective, the suppressor has to dominate the anode path impedance
to chassis. This means the suppressor inductance must be large compared to
anode-chassis path impedance. Short and wide anode leads, a low VHF impedance
plate tuning capacitor that is well-grounded to the main chassis, and compact
layout, work in concert to minimize required suppressor inductance.
The Most
Unstable Troublesome Tubes
The most
problematic
tubes for VHF
oscillation have
relatively large
elements and long
thin leads. Tubes of
this type have low
gain or are unusable
at VHF. This is because
elements in the tube
(shunt internal
capacitance combined with
series lead
inductances) are
actually resonant or connected to outside pins through high VHF impedances.
Internal
connecting leads
diameter and length are
almost always the major concern for parasitic instability.
Longer and
thinner internal
(and external) leads produce less stable and more difficult to use
tubes. Long and
thin leads move a tube element’s natural
self-resonance lower
in frequency and increase element
impedances. This
causes unwanted self-oscillation even with tiny
amounts of
anode-grid
feedback capacitance.
- A few examples
of common troublesome
tubes are
811A’s, 572B’s,
833’s, 4-1000A’s,
3CX1200A7’s, and
3CX1200D7’s. - A few tubes of moderate
instability are
3-500Z, 3-1000Z, and
4-400A’s. - Examples of tubes having
virtually
unconditional VHF
stability are the
3CX800A7, 3CX1200Z7,
3CX1500A7/8877,
3CX3000F7, and
3CX5000/3CPX5000/YU-156
series.
Looking at the
above, tubes with
thinnest and longest
leads are most
troublesome. These
tubes also provide
poorest intentional VHF
performance.
The most
troublesome tubes
listed above tend to
oscillate in the
lower-VHF range,
between 40 and 120
MHz. The typical
instability
frequency of an 811A
or 572B is around
80-100 MHz, assuming
grid leads are
short and direct to
the chassis.
The plot on left
is the feedthrough
loss of an 811A tube
in a shielded test fixture.
The tube can
oscillate on any
frequency where loss
is around -25 dB or
less.
Adding series capacitors from the grid to ground does not help, the result is
essentially no change. If the capacitor is too small, or the leads too long,
things get worse:
Note the 572B tube on left has very long, thin, grid leads. There are three inches
of very thin wire to get a chassis ground connection to the control grid! This
tube is typical of 572B and 811A triodes.
This tube was not designed with high frequencies in mind. For higher radio
frequencies, the second thinner socket pin could have provided an additional
parallel grid connection. The grid could have used much heavier and shorter
internal leads.
Moderately stable
tubes tend to
oscillate at
100-200MHz.
3-500Z’s, for
example, generally
are most unstable
from 150-200 MHz.
A very important
thing to remember! The closer a
tube’s instability
frequency is to the
operating frequency,
the more likely it
is to have tank-damaging
oscillations. This
is because the tube
might actually
oscillate on or very
near the tank
circuit’s resonant
frequency. It also
is much more
difficult to
stabilize a tube
with a low grid-frequency resonance
without severely
impacting desired
operating frequency
efficiency and gain.
Anode Circuit
Layout
Anode circuit
layout can
contribute to VHF
instability. Long
thin leads from the
tube anode connector
to the chassis at
VHF are a problem.
Problems can occur
when thin (and long)
plate blocking
capacitor leads,
thin and/or long
wiring, and poor
mounting of the
plate tuning
capacitor are used.
The anode path, from tube through blocking capacitor and through the plate
tuning capacitor to the chassis, is also an important
VHF path. This is true even
if the amplifier
only
intentionally
operates on HF.
To maximize
stability:
- Use wide anode
circuit leads
from the tube to
the tuning
capacitor. - Mount the
tuning capacitor
directly on the
chassis, or on a
large metallic
groundplane area
that is
thoroughly
bonded to the
chassis at many
points. - Use a
low-inductance
plate blocking
capacitor. - Keep all leads
as short as
possible, even
if it is at the
expense of
“looking
pretty”
with
perfectly
aligned
90-degree wiring angles. - Use the
chassis as a
groundplane and
as an input-to-output
shield. Keep the
tank circuit’s ground connection point common to the
grid ground
point in a
grounded grid
amplifier, but NEVER by using long leads. Use the chassis, or any large wide
groundplane, for this path. Not wires! - Don’t ground
tank capacitors
exclusively or
primarily to a
front panel. Ground them to the same metal as the grid, if possible.
Grid Circuit
Layout
The grid circuit
layout is probably
the single most
important area for
insuring a stable
design. Long thin
leads from the tube
grid connector to
chassis are a
problem at VHF.
Problems often occur
from physically thin and long
bodied
grid capacitors or thin
and/or needlessly
long grid wires or wiring.
The best idea is to ground
grids directly to
the chassis through
ground lugs mounted directly on the
chassis immediately
adjacent to
grid pins. Always
think “zero length
grid leads”!
To maximize
stability:
- Use wide
low-inductance
grid leads from
the tube socket
directly to the
chassis, connecting
grid grounding
leads to
the closest
possible point.
Ideally
use ground
lugs right at
the grid pins
(rather than
using socket
mounting screws)
for grounding. - Use low-pass
pi-network or
parallel tuned
networks as
input matching
circuits. - Mount any
swamping or grid load
resistors right
at or on
the tube socket
so leads are very
short. - Mount the
low-pass or
bandpass input
matching system
near the tube,
or use
exceptionally
low-impedance
transmission
lines to reach
the input
matching system. - Keep all grid
connections as
short as
possible, even
if it is at the
expense of
having wiring
“look
pretty”
with all
perfectly
aligned
90-degree angles.
What Does the
Parasitic Suppressor
Do?
The parasitic
suppressor normally
has two components
in parallel, a
resistor and an
inductor. At low
frequencies, the
path through the
inductor dominates
the system. At very
high frequencies,
the resistor
dominates the system
(assuming it is a
low-inductance
resistor).
One common
problem is people
assume brown carbon
resistors are
non-inductive. That
isn’t the case. For
an example, look at
the following
resistors:
I have non-inductive resistors suitable for VHF suppressors.
Old Parasitic System AL811H
The photo below shows the original standard parasitic suppression system used
in the AL811H amplifier:
All of the
spiral-conductor
resistors above have
significant
inductance at VHF,
and make very
ineffective
suppressors unless
the reactance is
cancelled. Only the
true carbon
composition
resistors are useful
in non-resonant
standard
suppressors.
This is a typical
suppressor system,
including inductance
of the anode lead:
In this case V1
represents the tube.
The following is a
simulation of
currents in the
suppressor:
Starting at
30MHz, the ratio of
current in the
inductor to current
in the resistor
is:
Frequency -I(L1)
-I(R1)
30MHz
0.0047
0.0015
60
0.0041
0.0026
90
0.0034
0.0034
120
0.0029
0.0037
160
0.0024
0.0041
190
0.0021
0.0042
220
0.0018
0.0043
This tells us
something very
important. The
INDUCTOR dominates
only at low
frequencies. At
30MHz, current in
the inductor is
three times current
in the resistor.
At 190MHz, in the
range of the
instability
frequency of a
3-500Z, the resistor
has twice the
current as the
inductor.
This tells us any
changes in INDUCTOR
design or inductor Q
(such as use of
nichrome wire)
mainly lowers low
frequency Q. It
would have virtually
no effect on very
high frequency Q of
the system.
- The dominant
factor in
controlling VHF
Q is the
resistor value,
and any
reactance in the
resistor path - The dominate
factor in
determining HF Q
and performance
is the inductor
value, and any
changes in
inductor Q
This has been my
point all along with
the Measure’s
nichrome suppressor.
Measures claims,
incorrectly, his
suppressors provide
lower VHF Q while,
in fact, they do
exactly the
opposite! A typical
Measures hairpin
suppressor actually
produced
significantly higher
system Q in the
anode of a 3-500Z
(nearly twice the
VHF Q), because the
equivalent Rp of the
suppressor in series
with the anode lead
was lower!
The reasons HF
PA’s arc are
explained at other
pages of this site,
and include
incorrect relay
sequencing, load
faults, as well as
improper tuning and
exciter transients.
Reducing VHF
Q
If we want a
lower VHF Q, while
maintaining high LF
Q and efficiency,
the system must
shift current into
the resistor faster
as frequency
increases. The
suppressor must also
have higher Rp, so
it dominates the
anode path
inductance that is
in series with the
suppressor.
While Measures
openly touts his
“low-Rp
suppressor”,
the fact is a low Rp
suppressor results
in higher anode
system Q!
A Truly Improved
Parasitic
Suppressor
In order to
improve VHF stability by reducing VHF Q and reduce VHF gain, we
must have a series resistance dominate
the anode system’s impedance at VHF.
This means, in a
frequency sweep
simulation, the
ratio of currents in
the resistance to
current in the
inductance must be
as high as possible.
Let’s call that
slope the rate
of transfer.
The rate of
transfer can be
increased by adding
a small value of
capacitance in
series with the
resistor:
The old
suppressor was:
Frequency -I(L1)
-I(R1)
Ratio
30MHz
0.0047
0.0015
3
60
0.0041
0.0026
1.6
90
0.0034
0.0034
1
120
0.0029
0.0037
.78
160
0.0024
0.0041
.58
190
0.0021
0.0042
.5
220
0.0018
0.0043
.42
The new one:
Frequency -I(L1)
-I(R1)
Ratio
30MHz
0.0069
0.0026
2.6
60
0.0050
0.0055
.9
90
0.0027
0.0052
.52
120
0.0019
0.0050
.38
160
0.0013
0.0048
.27
190
0.0011
0.0047
.23
220
0.0009
0.0047
.19
Graphically we
see the currents
are:
The green curve
is current through
the inductor, the
red curve shows
current through the
resistor. Notice how
flat current is in
the resistor, and
how sharp roll off
of current in the
inductor becomes.
This means we
will have very low anode
SYSTEM
Q starting at a low
VHF frequency of
50-60MHz, and
continuing up to
UHF.
Dissipation in the
resistor is still
reasonable at HF,
efficiency and tank
Q at the operating
frequency remain
high, yet VHF
suppression is
greatly improved.
Selecting
Component Values
Optimum resistor
value can
be determined by
network analyzer measurements, or
determined
empirically.
If the anode path
to chassis is long and thin,
the VHF impedance will
be very high. A high
anode path impedance
(thin or long leads)
requires higher
values of
resistance, because
we want the resistor
to dominate the
anode system
impedance. The best
value for a resistor
is generally one
that is
approximately equal
to, or slightly
higher than, the
anode path reactance
at the frequency of
instability. This ensures an upper-VHF Q approaching 1, and a broad dampening
bandwidth.
That impedance
can be measured on
an impedance test
set, or through other methods
by creative
engineers or
technicians. As
a general rule in good HF tank systems, long,
thin, anode leads
(i.e. 811A’s) require
100-150 ohms of
resistance while
shorter thicker
anode leads (i.e. 3-500Z) require 30-100
ohms of resistance.
Stable tubes with
external anodes
can often use
natural anode lead
resistance of
brass or other
materials, or even hairpins, to
adequately dampen
anode path
reactance. Exceptionally stable tubes with short internal grid leads exiting on
a grid ring often require no suppression at all, if the grid ring is grounded
directly to the chassis.
At the frequency of instability, the suppressor inductor must present
significantly higher
reactance than
suppression resistance
values. The high inductive reactance
causes the majority
of current to flow
through the
suppressor resistor at very high frequencies,
not through the
inductor.
Looking at
amplifier designs,
we will find tubes
like 811A’s
generally have
higher resistor values and many
turns in the
suppressor inductor. Tubes
like 3-500Z’s have
significantly fewer
turns, especially
when grid leads are
kept very short and
direct to the
chassis, and can use lower
value resistors.
- The less stable
the tube at low VHF,
the larger the
inductor must be. - The longer and thinner the tank leads to the tank input (variable)
capacitor and the longer the capacitor to chassis path, the higher
suppressor resistance and inductance must be.
One way to view
this is to consider
the frequency
response of a Hi-fi
amplifier. Larger
values of plate load
resistors in
amplifier stages
reduce
higher-frequency
gain. The same is
true in RF power amplifiers. Lower frequencies
of instability
require larger
inductors, so the RF
path is shifted over
to the resistor at a
lower frequency.
Uses For
Improved Suppressors
Series-resonant
suppressors are used
with slightly
inductive resistor
paths, and
larger-than- normal
shunt inductors. A
small capacitor is
placed in series
with the slightly inductive
resistor path, and
this capacitor
series-tunes the
resistor path to a very broad VHF resonance. This
results in a very
rapid shift of
current into the
resistor as
frequency is
increased. This
works well with
amplifiers operating
at 1/3 to 1/2 the
instability
frequency. It
minimizes resistor
heat on upper HF while providing
perfect lower or mid-VHF stability.
Typical
amateur applications for shunt L series R-C suppressors are
3CX1200A7 and D7
tubes, 572B tubes,
and 811A tubes.
Shunt suppressors
with series-resonant
tuning to ground are also
sometimes used, the
normal application
is very high power
stages with
substantial
anode-to-tank
currents. These
suppressors consist
of a series R/L/C
system, where the C
is normally just
stray capacitance to
the tube anode.
Sometimes these
suppressors take the
form of a ferrite
block placed between
the anode and
chassis. The
inductance of the
block series-tunes
stray capacitance,
and the losses act
like a damping
resistance in series
with the anode-to-chassis path. I’ve
stabilized 50-100kW
VHF transmitter
designs using shunt
suppression.
Other
Instability
Some PA systems
are prone to
oscillation at low
frequencies. Yaesu
and Dentron
amplifiers using
572B’s, and the
Collins amplifier
using 811A’s are
good examples of
production
amplifiers with
stability problems. These amplifiers
tend to oscillate
NEAR the operating
frequency. All of these
amplifiers, except
the Yaesu, use tubes
with high
anode-to-grid
feed-through
capacitance and no
neutralization.
Worse, the Collins
floats the grids for
RF, reducing the
already poor
isolation of
anode-to-cathode
feedback path in the
811A.
Yaesu uses one of
the poorest
engineered feedback
systems of all, with
a capacitor from the
output of the pi
section back to the
cathode! Phase shift
in that path would
vary wildly with
tank circuit tuning
and load impedance
on the PA, as would
the amount of
feedback!
The Yaesu
amplifier is a
particular problem
with Chinese 572B
tubes, because grid
mu is lower.
Negative grid bias
has LESS of an
effect on cathode
current, so the
Chinese (and
Russian) tubes draw
extra quiescent
current when the
antenna relay is
open. This
additional current
allows the tube to
amplify while the
amp is in standby.
Since the antenna
and input source are
removed in standby,
and the improperly
designed feedback
path to the tank
output remains in
place, the PA
oscillates near the operating frequency with no load! Voltage in
the tank builds up
to many thousands of
volts because energy is not extracted
to a load. The fact
the oscillation is
at a low frequency
allows the
bandswitch to see
the full voltage,
and it often fails.
Amplifiers can
create extremely
large voltages
when RF is applied
and a load is not
present!
All of the
amplifiers discussed
above would be
greatly improved by:
- Adding a
proper bridge
neutralization
circuit like
Heathkit,
Ameritron, and
Gonset used in
811 amplifiers. - Grounding the
grids either
directly or
through low
reactance
very-short-lead
capacitors,
directly between
the socket’s
grid pin and
chassis. - Using the
improved
suppressor
outlined above
to de-Q the amp
at lower VHF.
A Common “Bad Grid Idea” Super Cathode Drive
Floating grids on capacitors to add “negative feedback” is one of the worse things
every done in
grounded-grid
triode PA’s . This bad idea appears in the Collins
30L1 811A amplifier, and
Japanese
manufacturers copied
the bad idea into
their power
amplifiers. Heathkit
was also a victim of
this engineering
gaff in the SB-220 and SB-221 amplifiers. Here is how it started and filtered
through Ham gear:
When I was
designing PA’s in
the late 70’s and
early 80’s, an
employee of Eimac
(who was also an author of many
articles and a
popular Radio Handbook) put
considerable
pressure on me to
float the grids of
3-500Z PA’s
through small mica
capacitors. He
called the circuit a
“super-cathode
driven” amplifier.
He wrote letters and
called frequently,
asking why I would
not float the grids
through small mica
capacitors.
This
quite likable fellow
creatively
“borrowed” this idea
from the Collins
30S1, which was
actually a proper
application for this
type of system. This system works in the 30S1 because it is a
cathode-driven class
AB1 tetrode. The
30S1, unlike later “copy-cats” using the floating-grid circuit, has zero
control grid
current. The grid
has very high impedance all through the RF cycle. The high grid-cathode
impedance does not shunt the
upper capacitor
divider with the low
drive-varying grid
resistance of stages with control grid current.
Essentially R1 (see
the circuit below)
is infinite in the
Collins 30S1. The
30S1, unlike
triode copy cats, has a
directly grounded
screen. The screen
shields the RF input
(cathode) from the RF output (anode).
The theory seems pretty
simple on the
surface. Floating
control grids
through small mica
capacitors forms a
capacitive voltage
divider, with the
small
grid-to-ground
bypass capacitors forming
the grounded half of
a capacitive voltage divider. The
small
internal
cathode-to-grid
internal tube capacitance forms the upper leg
of this voltage
divider. Driving
power requirements are
increased by this
negative feedback
(the grid partially follows the RF
cathode voltage,
reducing effective
grid/cathode voltage
and reducing effective driving
power applied to the grid). In theory, the
amplifier should be
“cleaner” and, with
reduced power gain, be a
closer match to
higher power exciters.
The typical circuit is as follows:
Super cathode drive theory is the cathode to grid capacitance forms a divider with the grid bypass capacitor. This somewhat works in a class AB1 tetrode or pentode, because the cathode to grid circuit never biases into conduction. The idea falls apart with grid current in any amplifier, as well as in any triode.
After some
thought, experiments, and
questioning other engineers, I found no one actually measured
performance or
calculated feedback
over a wide range of
operating
frequencies and control grid
currents. It was assumed since everyone did it and an Eimac staffer endorsed it,
super-cathode was already confirmed technically sound.
Good Feedback Dividers
In a good
capacitive divider, sampled feedback
voltage would be constant
in both amplitude
and phase regardless
of frequency, power
levels, and tuning.
To be a “good” capacitive
divider, the
reactance of capacitors
C1 and C2 would
have to totally
dominate system
impedances. This is where the wheels fall off
“super cathode drive”.
The basic circuit
the Eimac marketing engineer and prominent handbook author promoted, and
that Heath and
others used, was
similar to this circuit:
The grid connects
at the junction of
C1 and C2, while the
cathode connects to
the top of C2.
C2 is the
internal stray G-K
capacitance of the
tube
R1 is the
time-varying grid
to cathode impedance
R2 is only added to
allow us to see the
input impedance
change of the
divider on the SPICE
model.
Sweeping the
system from 100KHz
to 30MHz shows
the following:
We find a huge
spike in
grid-to-ground
impedance at 2MHz,
and very uneven
response above that
range. By
manipulating the
value of L1 (the
grid chokes) we can
move the spike
around, but we are
ALWAYS left with
some low frequency where
the grid isn’t
grounded!
The Heathkit SB220, for example, peaks below
the 160-meter band.
This is a
very serious
violation of
good engineering
practices in any
grounded-grid PA,
and is actually at
the root of
VLF and HF stability problems
in a few popular PA’s.
Collins, for
example, had a
series of field
modifications to the
30L1 grid system.
They kept moving the spike around, trying to stabilize the amplifier. The best idea for
the 30L1 Collins would
have been to abandon
the silly notion
this system adds
stable controlled negative
feedback, and
change the amplifier
back to a true
grounded grid with neutralization. If
Collins wanted negative
feedback in the 30L1, the
PROPER
method would have
been the addition of
a
resistor in series
with the cathode
feed point near the
tubes! We never want to float the grids in a
grounded-grid triode amplifier.
There are
obviously several major
flaws with the
super-cathode drive
concept. Grid
current causes
grid-to-cathode
impedance to
constantly vary with
drive level.
When grid current is
absent, the
grid-to-cathode
impedance is nearly
an open circuit.
Grid-to-cathode
capacitance
dominates the upper
half of the divider,
and everything
appears to work as
planned. Unfortunately, a
problem appears
whenever the grid
draws current. Even
the tiniest amount
of grid current
causes
grid-to-cathode
impedance to
decreases rapidly.
With only a few
dozen milliamperes
of grid current,
grid impedance drops
to a few hundred
ohms or less. As
grid current is
drawn, the
decreasing grid
impedance dominates
the upper leg of the
voltage division
circuit!
There are also
new potentially
destabilizing
resonances added in
the grid path.
This system
causes four major
problems:
- Grid drive is
effectively
reduced as
operating
frequency
is increased.
This is the
opposite of what
we need! We need
more drive to
offset system
inefficiencies on
higher
frequencies. - Feedback
starts to show
significant
phase-lag with
increased drive,
especially on
lower bands. - Grid-to-chassis
impedance at
VHF and LF is
increased,
making the
amplifier much less
stable. An
SB-220 heath
amplifier for
example required
nearly twice the
parasitic choke
inductance when
the “super
cathode” circuit
was used. Still,
because of
pressure from this
person, the
circuit was added! - Protection for the exciter and cathode system, in the event of a tube
arc, is greatly reduced. (see the mods on
this link for the 572B and 811H amplfiers)
When I tested
several amplifiers
with this alleged
“super-cathode”
system added, IMD
performance
became significantly worse under
some operating
conditions.
Stability also
significantly
decreased. Several
amplifiers I tested
using 572B, 3-1000Z,
and 3-500Z tubes all
had higher
intermodulation
distortion and
required larger
parasitic chokes
when this
super-cathode system
was added!
Unless you have a
class AB1 tetrode or
pentode, ground the
control grids
directly with
short heavy leads or
use low-inductance
high-value capacitors
with very short
leads to ground the control grids! The “super cathode drive” system system
does not belong in any grounded grid triode amplifier. Get rid of it!
Summary
I hope this
information is
useful, and helps
people understand
what really goes on
in a parasitic
suppression system.
As time permits, I
add more articles
about curing unique
problems in
amplifiers, and
diagnosing amplifier
failures. I hope
these pages are a
good start.
Please pass this
web address along to
others.
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